Signal processing apparatus, signal processing method, and signal processing program

ABSTRACT

To obtain a high-quality enhanced signal, there is provided an apparatus including a transformer that transforms a mixed signal, in which a first signal and a second signal coexist, into a phase component for each frequency and one of an amplitude component and a power component for each frequency, a change amount generator that generates a change amount of the phase component at a predetermined frequency by using a series of data with a cross-correlation weaker than that of the phase components and randomness lower than that of random numbers, a phase controller that controls the phase component by using the change amount provided from the change amount generator, and an inverse transformer that generates an enhanced signal by using the phase component having undergone control processing by the phase controller.

TECHNICAL FIELD

The present invention relates to a signal processing technique ofcontrolling the phase component of a signal.

BACKGROUND ART

As examples of a technique of performing signal processing bycontrolling the phase component of a signal, patent literature 1 andnon-patent literature 1 disclose noise suppression techniques which payattention to a phase spectrum. In the techniques described in patentliterature 1 and non-patent literature 1, an amplitude spectrumpertaining to noise is suppressed, and at the same time, the phasespectrum is shifted by a random value of up to π/4. The techniquesdescribed in patent literature 1 and non-patent literature 1 implement,by shifting the phase spectrum at random, suppression of noise whichcannot be suppressed by only attenuation of the noise spectrum.

CITATION LIST Patent Literature

-   Patent literature 1: International Publication No. 2007/029536-   Non-patent literature 1: Akihiko Sugiyama, “Single-Channel    Impact-Noise Suppression with No Auxiliary Information for Its    Detection,” Proc. IEEE Workshop on Appl. of Sig. Proc. to Audio and    Acoustics(WASPAA), pp. 127-130, Oct. 2007.

SUMMARY OF THE INVENTION Technical Problem

However, as in patent literature 1 and non-patent literature 1, to shiftthe phase spectrum at random, it is necessary to generate a randomnumber. As a result, a calculation amount for generating a random numberis added.

The present invention enable to provide a signal processing technique ofsolving the above-described problem.

Solution to Problem

One aspect of the present invention provides an apparatus characterizedby comprising:

a transformer that transforms a mixed signal, in which a first signaland a second signal coexist, into a phase component for each frequencyand one of an amplitude component and a power component for eachfrequency;

a change amount generator that generates a change amount of the phasecomponent at a predetermined frequency by using a series of data with across-correlation weaker than that of the phase components andrandomness lower than that of random numbers;

a phase controller that controls the phase component by using the changeamount provided from the change amount generator; and

an inverse transformer that generates an enhanced signal by using thephase component having undergone control processing by the phasecontroller.

Another aspect of the present invention provides a method characterizedby comprising:

transforming a mixed signal, in which a first signal and a second signalcoexist, into a phase component for each frequency and one of anamplitude component and a power component for each frequency;

generating a change amount of the phase component at a predeterminedfrequency by using a series of data with a cross-correlation weaker thanthat of the phase components and randomness lower than that of randomnumbers;

controlling the phase component by using the change amount generated inthe generating the change amount; and

generating an enhanced signal by using the phase component havingundergone control processing in the controlling.

Still other aspect of the present invention provides a program forcausing a computer to execute a method, characterized by comprising:

transforming a mixed signal, in which a first signal and a second signalcoexist, into a phase component for each frequency and one of anamplitude component and a power component for each frequency;

generating a change amount of the phase component at a predeterminedfrequency by using a series of data with a cross-correlation weaker thanthat of the phase components and randomness lower than that of randomnumbers;

controlling the phase component by using the change amount generated inthe generating the change amount; and

generating an enhanced signal by using the phase component havingundergone control processing in the controlling.

Advantageous Effects of Invention

According to the present invention, it is possible to provide a signalprocessing technique of controlling the phase component of an inputsignal without generating any random number.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a block diagram showing the schematic arrangement of a signalprocessing apparatus according to the first embodiment of the presentinvention;

FIG. 2 is a block diagram showing the schematic arrangement of a noisesuppression apparatus according to the second embodiment of the presentinvention;

FIG. 3 is a block diagram showing the arrangement of a transformerincluded in the second embodiment of the present invention;

FIG. 4 is a block diagram showing the arrangement of an inversetransformer included in the second embodiment of the present invention;

FIG. 5 is a block diagram showing the schematic arrangement of a noisesuppression apparatus according to the third embodiment of the presentinvention;

FIG. 6 is a block diagram showing the schematic arrangement of a changeamount generator included in the noise suppression apparatus accordingto the third embodiment of the present invention;

FIG. 7 is a block diagram showing the schematic arrangement of a noisesuppression apparatus according to the fourth embodiment of the presentinvention;

FIG. 8 is a block diagram showing the schematic arrangement of anamplitude controller 708 included in the noise suppression apparatusaccording to the fourth embodiment of the present invention;

FIG. 9 is a view showing a signal flow when no phase rotation isperformed in the frequency domain according to the fourth embodiment ofthe present invention;

FIG. 10 is a view showing a signal flow when phase rotation is performedin the frequency domain according to the fourth embodiment of thepresent invention;

FIG. 11 is a timing chart showing overlap addition of frames when nophase rotation is performed in the frequency domain according to thefourth embodiment of the present invention;

FIG. 12 is a timing chart showing overlap addition of frames when phaserotation is performed in the frequency domain according to the fourthembodiment of the present invention;

FIG. 13 is a view showing the vector of a frequency domain signal whenphase rotation is performed in the frequency domain according to thefourth embodiment of the present invention;

FIG. 14 is a view showing the vector of a frequency domain signal whenno phase rotation is performed in the frequency domain according to thefourth embodiment of the present invention;

FIG. 15 is a block diagram showing the arrangement of a noisesuppression apparatus according to the fifth embodiment of the presentinvention;

FIG. 16 is a block diagram showing the arrangement of a noisesuppression apparatus according to the sixth embodiment of the presentinvention;

FIG. 17 is a block diagram showing the arrangements of a phasecontroller and amplitude controller according to the seventh embodimentof the present invention;

FIG. 18 is a block diagram showing the arrangements of a phasecontroller and amplitude controller according to the eighth embodimentof the present invention;

FIG. 19 is a block diagram showing the arrangements of a phasecontroller and amplitude controller according to the ninth embodiment ofthe present invention;

FIG. 20 is a block diagram showing the schematic arrangement of a noisesuppression apparatus according to the 10th embodiment of the presentinvention;

FIG. 21 is a block diagram showing the arrangements of a phasecontroller and amplitude controller according to the 10th embodiment ofthe present invention;

FIG. 22 is a block diagram showing the schematic arrangement of a noisesuppression apparatus according to the 11th embodiment of the presentinvention; and

FIG. 23 is a block diagram showing the schematic arrangement of a noisesuppression apparatus according to another embodiment of the presentinvention.

DESCRIPTION OF THE EMBODIMENTS

Preferred embodiments of the present invention will now be described indetail with reference to the drawings. It should be noted that therelative arrangement of the components, the numerical expressions andnumerical values set forth in these embodiments do not limit the scopeof the present invention unless it is specifically stated otherwise.

First Embodiment

FIG. 1 is a block diagram showing the schematic arrangement of a signalprocessing apparatus 100 according to the first embodiment of thepresent invention. Referring to FIG. 1, the signal processing apparatus100 includes a transformer 101, a phase controller 102, a change amountgenerator 103, and an inverse transformer 104.

The transformer 101 transforms a mixed signal 110, in which the firstand second signals coexist, into a phase component 120 for eachfrequency and an amplitude component or power component 130 for eachfrequency.

The change amount generator 103 generates a change amount of a phasecomponent at a predetermined frequency by using a series of data with across-correlation weaker than that of the phase components 120 andrandomness lower than that of random numbers. The phase controller 102controls the phase component 120 by using the change amount providedfrom the change amount generator 103. The inverse transformer 104generates an enhanced signal 170 using a phase component 140 havingundergone control processing by the phase controller 102.

With the above arrangement, it is possible to control the phasecomponent 120 using a series of data with a cross-correlation weakerthan that of the phase components 120 and randomness lower than that ofrandom numbers, thereby efficiently implementing suppression of noisewhich cannot be suppressed by only attenuation of the amplitudespectrum.

Second Embodiment Overall Arrangement

A noise suppression apparatus 200 according to the second embodiment ofthe present invention will be described with reference to FIGS. 2 to 4.FIG. 2 is a block diagram showing the overall arrangement of the noisesuppression apparatus 200. The noise suppression apparatus 200 accordingto this embodiment functions as part of an apparatus such as a digitalcamera, notebook personal computer, or mobile phone. However, thepresent invention is not limited to this. The noise suppressionapparatus 200 is applicable to all information processing apparatusesrequested to remove noise from an input signal. When, for example, anoperation such as pressing of a button is performed near a microphone,the noise suppression apparatus according to this embodimentappropriately removes an impulsive sound generated by the operation ofthe button. In brief, an impulsive sound is appropriately removed bytransforming a signal including the impulsive sound into a frequencydomain signal, and controlling the phase component in the frequencydomain using a series of data with a weak cross-correlation.

A deteriorated signal (a signal in which a desired signal and noisecoexist) is supplied as a series of sample values to an input terminal206. When a deteriorated signal is supplied to the input terminal 206, atransformer 201 performs transform such as Fourier transform for thesupplied deteriorated signal, and divides the resultant signal into aplurality of frequency components. The transformer 201 independentlyprocesses the plurality of frequency components for each frequency. Thefollowing description pays attention to a specific frequency component.The transformer 201 supplies a deteriorated signal amplitude spectrum(amplitude component) 230 of the plurality of frequency components to aninverse transformer 204. The transformer 201 supplies a phase spectrum(phase component) 220 of the plurality of frequency components to aphase controller 202 and a change amount generator 203. Note thatalthough the transformer 201 supplies the deteriorated signal amplitudespectrum 230 to the inverse transformer 204, the present invention isnot limited to this. The transformer 201 may supply a power spectrumcorresponding to the square of the deteriorated signal amplitudespectrum 230 to the inverse transformer 204.

The change amount generator 203 generates a change amount by using thedeteriorated signal phase spectrum 220 received from the transformer201, and supplies the change amount to the phase controller 202. The“change amount” of the phase is a concept including the “rotationamount” and “replacement amount” of the phase, and indicates the controlamount of the phase. The phase controller 202 reduces the phasecorrelation by changing the deteriorated signal phase spectrum 220supplied from the transformer 201 by using the change amount suppliedfrom the change amount generator 203, and supplies the resultant data asan enhanced signal phase spectrum 240 to the inverse transformer 204.The inverse transformer 204 performs inverse transform by composing theenhanced signal phase spectrum 240 supplied from the phase controller202 and the deteriorated signal amplitude spectrum 230 supplied from thetransformer 201, and supplies the result of inverse transform as anenhanced signal 270 to an output terminal 207.

<<Arrangement of Transformer>>

FIG. 3 is a block diagram showing the arrangement of the transformer201. As shown in FIG. 3, the transformer 201 includes a frame divider301, a windowing unit 302, and a Fourier transformer 303. A deterioratedsignal sample is supplied to the frame divider 301, and divided intoframes for every K/2 samples where K is an even number. The deterioratedsignal sample divided into frames is supplied to the windowing unit 302,and multiplied by a window function w(t). A signal obtained byperforming windowing for an input signal y_(n)(t) (t=0, 1, . . . ,K/2−1) of the nth frame by w(t) is given by:

y _(n)(t)=w(t)y _(n)(t)  (1)

The windowing unit 302 may partially superimpose (overlap) and windowtwo successive frames. Assuming that the overlapping length is 50% ofthe frame length, the windowing unit 302 outputs the left-hand side ofequation (2) below obtained for t=0, 1, . . . , K/2−1.

y _(n)(t)=w(t)y _(n-1)(t+k/2)

y _(n)(t+K/2)=w(t+K/2)y _(n)(t)  (2)

For a real signal, the windowing unit 302 may use a symmetric windowfunction. The window function is designed so that, when the phasecontroller 202 performs no control operation, the input signal of thetransformer 201 and the output signal of the inverse transformer 204coincide with each other by excluding a calculation error. This meansw(t)+w(t+K/2)=1.

The following description will continue by exemplifying a case in whichtwo successive frames are made to overlap each other by 50% andwindowed. For example, the windowing unit 302 may use, as w(t), aHanning window given by:

$\begin{matrix}{{w(t)} = \left\{ \begin{matrix}{{0.5 + {0.5{\cos \left( \frac{\pi \left( {t - {K/2}} \right)}{K/2} \right)}}},} & {0 \leq t < K} \\{0,} & {otherwise}\end{matrix} \right.} & (3)\end{matrix}$

In addition, various window functions such as a Hamming window andtriangle window are known. The windowed output is supplied to theFourier transformer 303, and transformed into a deteriorated signalspectrum Y_(n)(k). The deteriorated signal spectrum Y_(n)(k) isseparated into a phase and amplitude. A deteriorated signal phasespectrum arg Y_(n)(k) is supplied to the phase controller 202 and changeamount generator 203, and a deteriorated signal amplitude spectrum|Y_(n)(k)| is supplied to the inverse transformer 204. As describedabove, the power spectrum may be used instead of the amplitude spectrum.

<<Arrangement of Inverse Transformer>>

FIG. 4 is a block diagram showing the arrangement of the inversetransformer 204. As shown in FIG. 4, the inverse transformer 204includes an inverse Fourier transformer 401, a windowing unit 402, and aframe composition unit 403. The inverse Fourier transformer 401 obtainsan enhanced signal (the left-hand side of equation (4) below) bymultiplying the deteriorated signal amplitude spectrum 230 (|X_(n)(k)|)supplied from the transformer 201 and the enhanced signal phase spectrum240 (arg X_(n)(k)) supplied from the phase controller 202.

X _(n)(k)=|X _(n)(k)|·arg X _(n)(k)  (4)

The inverse Fourier transformer 401 performs inverse Fourier transformfor the obtained enhanced signal. The enhanced signal having undergoneinverse Fourier transform is supplied to the windowing unit 402 as aseries x_(n)(t) (t=0, 1, . . . , K−1) of time domain sample values inwhich one frame includes K samples, and multiplied by the windowfunction w(t). A signal obtained by performing windowing for the inputsignal x_(n)(t) (t=0, 1, . . . , K/2−1) of the nth frame by w(t) isgiven by the left-hand side of:

x _(n)(t)=w(t)x _(n)(t)  (5)

The windowing unit 402 may partially superimpose (overlap) and windowtwo successive frames. Assuming that the overlapping length is 50% ofthe frame length, the windowing unit 402 outputs and transmits, to theframe composition unit 403, the left-hand side of the following equationfor t=0, 1, . . . , K/2−1.

$\begin{matrix}\left. \begin{matrix}{{{\overset{\_}{x}}_{n}(t)} = {{w(t)}{x_{n - 1}\left( {t + {K/2}} \right)}}} \\{{{\overset{\_}{x}}_{n}\left( {t + {K/2}} \right)} = {{w\left( {t + {K/2}} \right)}{x_{n}(t)}}}\end{matrix} \right\} & (6)\end{matrix}$

The frame composition unit 403 extracts outputs of two adjacent framesfrom the windowing unit 402 for every K/2 samples, superimposes them,and obtains an output signal (the left-hand side of equation (7) below)for t=0, 1, . . . , K−1.

{circumflex over (x)} _(n)(t)= x _(n-1)(t+K/2)+ x _(n)(t)  (7)

The frame composition unit 403 transmits the obtained output signal tothe output terminal 207.

In FIGS. 3 and 4, the transform processes in the transformer 201 andinverse transformer 204 have been described as Fourier transformprocesses. However, another transform such as Hadamard transform, Haartransform, or Wavelet transform may be used, instead of Fouriertransform. When the transformer 201 and inverse transformer 204 use Haartransform, the need for multiplication is eliminated, and it is thuspossible to decrease the area on an LSI circuit. When the transformer201 and inverse transformer 204 use Wavelet transform, it is possible tochange the time resolution depending on the frequency, and thus animprovement in noise suppression effect can be expected.

After a plurality of frequency components obtained by the transformer201 are integrated, the change amount generator 203 may generate achange amount and the phase controller 202 may control the phase. Atthis time, it is possible to obtain higher sound quality by integratinga larger number of frequency components from a low-frequency domainwhere the discrimination ability of auditory properties is high toward ahigh-frequency domain where the discrimination ability is poor so thatthe bandwidth after integration becomes wide. When phase control isexecuted after integrating a plurality of frequency components, thenumber of frequency components to which phase control is applieddecreases, and the total amount of calculation can be reduced.

<<Operation of Change Amount Generator 203>>

The change amount generator 203 is supplied with the deteriorated signalphase spectrum 220 from the transformer 201, and generates a changeamount to reduce the phase correlation. Since the deteriorated signalphase spectrum 220 supplied from the transformer 201 is represented byarg Y_(n)(k) (0≦k<K), the change amount generator 203 can obtain anenhanced signal phase spectrum arg X_(n)(k) for which correlation hasbeen reduced, as given by:

arg X _(n)(k)=(−1)^(k) arg Y _(n)(k)  (8)

This corresponds to alternately inverting the signs of the phases.Instead of alternately inverting the signs, inversion may be performedfor every arbitrary integer smaller than K, as a matter of course.

The change amount generator 203 obtains a rotation amount Δarg Y_(n)(k)as a change amount necessary for phase control indicated by equation(8), as given by:

Δarg Y _(n)(k)={(−1)^(k)−1}arg Y _(n)(k)  (9)

That is, the change amount generator 203 generates the rotation amountΔarg Y_(n)(k) indicated by equation (9) as a change amount. Also, it ispossible to use:

Δarg Y _(n)(k)=arg Y _(n)(mod [k+K/2−1,K])  (10)

where mod [k, K] represents a remainder obtained by dividing k by K. Therotation amount Δarg Y_(n)(k) at this time corresponds to a phaseobtained by shifting the original phase by K/2 samples. It is apparentthat he shift amount is not limited to K/2, and may be an arbitraryinteger.

Alternatively, a phase at a position symmetrical to the position of theoriginal phase with respect to K/2 is set as the rotation amount ΔargY_(n)(k). This uses:

Δarg Y _(n)(k)=arg Y _(n)(mod [K−k+1,K])  (11)

Furthermore, it is possible to generate a change amount by combiningthese two kinds of processes, that is, sign inversion and addition ofthe shifted phase. That is,

Δarg Y _(n)(k)={(−1)^(k)−1}arg Y _(n)(mod [k+K/2−1,K])  (12)

or

Δarg Y _(n)(k)={(−1)^(mod(k+N/2−1,N))−1}arg Y _(n)(mod[k+K/2−1,K])  (13)

As for the shift addition, the shift amount K/2 can be changed. Forexample, if a frame number n at that time is set as the shift amount,the shift amount automatically changes with time. Similarly, inequations (12) and (13), the equation (10) may be combined instead ofequation (11).

Furthermore, constant multiplication can be combined with the selectivesign inversion of the phase and shift addition processing. For example,combining constant multiplication with equation (10) yields:

Δarg Y _(n)(k)=k·arg Y _(n)(mod [k+K/2−1,K])  (14)

This is an example of performing constant multiplication for a term toundergo shift addition by k corresponding to the position of the term.

Furthermore, it is possible to replace a plurality of phase samples. Forexample, k (0≦k<K/2) can be alternately applied with:

Δarg Y _(n)(k)=−arg Y _(n)(k)+arg Y _(n)(mod [K−k+1,K])

Δarg Y _(n)(mod [K−k+1,K])=−arg Y _(n)(mod [K−k+1,K])+arg Y_(n)(k)  (15)

An arbitrary integer smaller than K may be used, instead of 1.

Selective sign inversion of the phase, shift addition, constantmultiplication, and replacement have been described above. Theseprocesses can be selectively applied in accordance with the value of argY_(n)(k). For example, it is possible to apply the above-describedprocesses only when the value of arg Y_(n)(k) takes a positive value.Exemplifying the processing indicated by equation (10) yields:

$\begin{matrix}{{\Delta \; \arg \; {Y_{n}(k)}} = {{\frac{{{sgn}\left( {\arg \; {Y_{n}(k)}} \right)} + 1}{2} \cdot \arg}\; {Y_{n}\left( {{mod}\left\lbrack {{k + {K/2} - 1},K} \right\rbrack} \right)}}} & (16)\end{matrix}$

where sgn(•) represents an operator for extracting a sign. A fraction onthe right-hand side becomes 1 only when the phase takes a positivevalue, and becomes zero otherwise. It is therefore possible toselectively apply the processes in accordance with the value of argY_(n)(k). The correlation elimination processes using the change amountare different in the degree of correlation elimination and the necessarycalculation amount. In actual application, in consideration of thedegree of correlation elimination and the necessary calculation amount,appropriate processing is selected and used or the processes are used incombination.

As another correlation elimination method, there is provided a method ofobtaining the correlation of the phase samples arg Y_(n)(k) andeliminating the obtained correlation. For example, consider a case inwhich arg Y_(n)(k) is represented by linear combination of N−1 adjacentsamples. This establishes:

$\begin{matrix}{{\arg \; {Y_{n}(k)}} = {{\sum\limits_{j = {{mod}({k - K + {1 \cdot K}}\}}}^{k - 1}{a_{j}\arg \; {Y_{n}(j)}}} + {\delta_{L}(k)}}} & (17)\end{matrix}$

Alternatively, paying attention to the correlation in the oppositedirection can yield:

$\begin{matrix}{{\arg \; {Y_{n}(k)}} = {{\sum\limits_{j = {k + 1}}^{{mod}{({{k + K - 1},K})}}{a_{j}\arg \; {Y_{n}(j)}}} + {\delta_{R}(k)}}} & (18)\end{matrix}$

Note that δ_(L)(k) and δ_(R)(k) represent uncorrelated components(components with no correlation).

Modifying arg Y_(n)(k) using the relationship yields:

$\begin{matrix}{{{{\Delta arg}\; {Y_{n}(k)}} = {- {\sum\limits_{j = {{mod}{({{k - K + 1},K})}}}^{k}{a_{j}\arg \; {Y_{n}(j)}}}}}{or}} & (19) \\{{{\Delta arg}\; {Y_{n}(k)}} = {- {\sum\limits_{j = k}^{{mod}{({{k + K - 1},K})}}{a_{j}\arg \; {Y_{n}(j)}}}}} & (20)\end{matrix}$

In the above equations, it is not necessary to use all nonzero valuesa_(j). By using some values a_(j), it is possible to reduce thecalculation amount.

Although the correlation elimination effect decreases, it is possible tominimize a decrease in effect by selectively using large values a_(j).As an example, by using only the largest value a_(j), phase correlationelimination is performed based on:

Δarg Y _(n)(k)=−a _(jmax) arg Y _(n)(jmax)  (21)

where jmax represents the value of j with which a correlationcoefficient a takes a largest value. As compared with correlationelimination using N samples, it is possible to reduce the calculationamount necessary for correlation elimination.

The coefficient a_(j) in the above linear correlation equations is knownas a linear prediction coefficient (LPC) in voice encoding. It ispossible to obtain the LPC at high speed by using a Levinson-Durbinrecursion method. Also, it is possible to obtain the LPB using acoefficient update algorithm for an adaptive filter represented by anormalized LMS algorithm by using the difference (error) between theoriginal phase sample value and the prediction result.

The correlation may be eliminated by assuming linear combination ofK_(j)−1 samples (K_(j)<K), instead of linear combination of K−1 adjacentsamples. By decreasing the number of samples assumed for linearcombination in this way, it is possible to reduce the calculation amountnecessary for correlation elimination.

A case in which arg Y_(n)(k) is represented by linear combination of K−1adjacent samples has been exemplified. Similarly, a case in which argY_(n)(k) is represented by nonlinear combination of K−1 samples ispossible. That is, this establishes:

arg Y _(n)(k)=f _(NL) [arg Y _(n)(j)]|_(0≦j<K,j≠k)+δ(k)  (22)

where f_(NL)[•] represents a nonlinear function, and δ(k) represents anuncorrelated component. In this case, the change amount used forcorrelation elimination can be obtained by:

Δarg Y _(n)(k)=−f _(NL) [arg Y _(n)(j)]|_(0≦j<K,j≠k)  (23)

Correlation elimination using the nonlinear function can sufficientlyeliminate the correlation when data have a nonlinear correlation.

The nonlinear function can be generally approximated by a polynomial.When approximating the nonlinear function f_(NL)[•] by a polynomial ofarg Y_(n)(j), the kinds of arg Y_(n)(j) are limited, and its order canalso be limited. If, for example, only arg Y_(n)(k), arg Y_(n)(k+1), andthe squares of them are used, f_(NL)[•] is approximated by only the fourkinds of terms including arg Y_(n)(k), arg Y_(n)(k+1), and the squaresof them. Approximation of the nonlinear function can reduce thecalculation amount necessary for correlation elimination.

<<Operation of Phase Controller 202>>

The phase controller 202 obtains the enhanced signal phase spectrum 240arg X_(n)(k) by adding the change amount Δarg Y_(n)(k) supplied from thechange amount generator 203 to the deteriorated signal phase spectrum220 supplied from the transformer 201, and supplies the obtainedenhanced signal phase spectrum 240 to the inverse transformer 204. Thatis, this executes:

arg X _(n)(k)=arg Y _(n)(k)+Δarg Y _(n)(k)  (24)

The phase controller 202 can obtain the enhanced signal phase spectrum240 arg X_(n)(k) by replacing the change amount Δarg Y_(n)(k) suppliedfrom the change amount generator 203 with the deteriorated signal phasespectrum 220 supplied from the transformer 201 without adding the changeamount to the deteriorated signal phase spectrum 220, and supply theenhanced signal phase spectrum 240 to the inverse transformer 204. Thatis, the phase rotation amount equals the replacement amount of the phaseby executing:

arg X _(n)(k)=arg Y _(n)(k)−arg Y _(n)(k)+Δarg Y _(n)(k)  (25)

Note that although replacement is implemented by subtracting theenhanced signal phase spectrum itself, and adding the rotation amount inthis example, replacement may be implemented by simply replacing phasedata with the replacement amount.

As described above, the shape of the deteriorated signal phase spectrum220 is changed when the phase controller 202 changes the value of ΔargY_(n)(k) by using the change amount Δarg Y_(n)(k) generated by thechange amount generator 203. The change of the shape weakens thecorrelation of the deteriorated signal phase spectrum 220, therebyweakening the feature of the input signal.

Note that it is also possible to apply phase unwrapping prior to thephase processing described above. This is because the deterioratedsignal phase spectrum 220 has a range of ±π as a value range. That is,phase unwrapping is performed not to limit the value range to the rangeof ±π. Performing phase unwrapping makes it possible to obtain thecorrelation indicated by equation (15), (16), or (20) at high accuracy.Various methods can be applied for phase unwrapping, as described in B.Rad and T. Virtanen, “Phase spectrum prediction of audio signals,” Proc.ISCCSP2012, CD-ROM, May 2012 (non-patent literature 2) and S. T. Kaplanand T. J. Ulrych, “Phase Unwrapping: A review of methods and a noveltechnique,” Proc. 2007 CSPG CSEG Conv. pp. 534-537, May 2007 (non-patentliterature 3).

Third Embodiment Overall Arrangement

A noise suppression apparatus 500 according to the third embodiment ofthe present invention will be described with reference to FIG. 5. FIG. 5is a block diagram showing the overall arrangement of the noisesuppression apparatus 500. The noise suppression apparatus 500 accordingto this embodiment is the same as the noise suppression apparatus 200according to the second embodiment except for a change amount generator503. Only the change amount generator 503 will be explained and adetailed description of the remaining components will be omitted.

<<Arrangement of Change Amount Generator 503>>

FIG. 6 is a block diagram showing the arrangements of a phase controller202 and the change amount generator 503. As shown in FIG. 6, the changeamount generator 503 includes an amplitude holding unit 601 and anamplitude analyzer 602. The amplitude holding unit 601 holds adeteriorated signal amplitude spectrum 230, and supplies it to theamplitude analyzer 602.

A transformer 201 supplies a deteriorated signal phase spectrum 220 tothe phase controller 202, and the change amount generator 503 supplies aphase rotation amount to the phase controller 202. The phase controller202 rotates (shifts) the deteriorated signal phase spectrum 220 by therotation amount supplied from the change amount generator 503, andsupplies the rotation result as an enhanced signal phase spectrum 240 toan inverse transformer 204.

<<Change Amount Calculation 1 Using Amplitude>>

For example, the amplitude analyzer 602 sets, as a rotation amount, aproduct obtained by multiplying the deteriorated signal amplitudespectrum held in the amplitude holding unit 601 by π. Alternatively,even if the deteriorated signal amplitude spectrum held in the amplitudeholding unit 601 is collected in the frequency direction or time axisdirection, and directly set as a rotation amount, the same effects areobtained. The phase controller 202 changes (rotates or replaces) thedeteriorated signal phase spectrum at each frequency by using the changeamount generated by the change amount generator 503 based on thedeteriorated signal amplitude spectrum. Under the control of the phasecontroller 202, the shape of the deteriorated signal phase spectrum 220changes. The change of the shape can weaken the feature of noise.

<<Change Amount Calculation 2 Using Amplitude>>

The amplitude analyzer 602 may supply, as a rotation amount, the resultof normalizing the deteriorated signal amplitude spectrum 230 held inthe amplitude holding unit 601 to the phase controller 202. In thiscase, the amplitude analyzer 602 first obtains the average of thedeteriorated signal amplitude spectra 230 (K positive values). Theamplitude analyzer 602 obtains a product by multiplying, by π, aquotient obtained by dividing the deteriorated signal amplitude spectrum230 by the obtained average value, and sets the product as a rotationamount. Note that if the quotient is directly set as a rotation amountwithout multiplying the quotient by π, the similar effects are obtained.Since a variance can be made large with respect to a case in which nonormalization is performed, the correlation elimination effect for arotated phase can be enhanced. Also, the average can be obtained afterexcluding a value (outlier) extremely different from the remainingvalues. This can eliminate the adverse effect of the outlier, therebyobtaining a more effective rotation amount.

<<Change Amount Calculation 3 Using Amplitude>>

The amplitude analyzer 602 can normalize the distribution of thedeteriorated signal amplitude spectra 230, and then set a rotationamount. First, the amplitude analyzer 602 obtains a maximum value|X_(n)(K)|_(max) and a minimum value |X_(n)(K)|_(min) of thedeteriorated signal amplitude spectra 230 (K positive values). Theamplitude analyzer 602 subtracts the minimum value from the deterioratedsignal amplitude spectrum, and divides the subtraction result by thedifference between the maximum value and the minimum value. A productobtained by multiplying the obtained quotient by π is set as a rotationamount. That is, a rotation amount Δarg Y_(n)(k) is obtained by:

$\begin{matrix}{{\Delta \; \arg \; {Y_{n}(k)}} = {\frac{{{X_{n}(k)}_{n}} - {{X_{n}(k)}}_{\min}}{{{X_{n}(k)}}_{\max} - {{X_{n}(k)}}_{\min}} \cdot \pi}} & (26)\end{matrix}$

By obtaining the rotation amounts in this way, the rotation amounts aredistributed between 0 and π. It is, therefore, possible to enhance thecorrelation elimination effect for a rotated phase. Note that even ifthe quotient is directly set as a rotation amount without multiplyingthe quotient by π, the similar effects are obtained.

<<Change Amount Calculation 4>>

A change amount generator 503 can normalize the distribution of thedeteriorated signal amplitude spectra by an envelope, and set thenormalization result as a rotation amount. As for the envelope, forexample, a regression curve of the deteriorated signal amplitudespectrum is obtained based on N samples, and each sample is divided bythe value of the regression curve. The regression curve may be obtainedby using some of the N samples, or can be obtained by excluding anoutlier. By excluding an outlier, it is possible to eliminate theadverse effect of the outlier, thereby obtaining a more effectiverotation amount. The thus obtained quotients are distributed centered on1.

By applying normalization of the maximum value and minimum valuedescribed using equation (26) to the quotient, the rotation amount ΔargY_(n)(k) is obtained by:

$\begin{matrix}{{{\Delta arg}\; {Y_{n}(k)}} = {\frac{{{{\overset{\sim}{X}}_{n}(k)}_{k}} - {{X_{n}(k)}}_{\min}}{{{X_{n}(k)}}_{\max} - {{X_{n}(k)}}_{\min}} \cdot \pi}} & (27)\end{matrix}$

where |{tilde over (X)}_(n)(k)| represents the deteriorated signalamplitude spectrum normalized by the envelope. By obtaining the rotationamounts, the rotation amounts are uniformly distributed between π and−π, thereby enhancing the correlation elimination effect. Note that evenif the quotient is directly set as a rotation amount without multiplyingthe quotient by π, the similar effects are obtained.

Fourth Embodiment

The fourth embodiment of the present invention will be described withreference to FIG. 7. A noise suppression apparatus 700 according to thisembodiment is different from the second embodiment in that an amplitudecontroller 708 is used to compensate for a drop of the output levelcaused by phase control of a phase controller 202. The remainingcomponents and operations are the same as those in the second embodimentand a description thereof will be omitted.

As shown in FIG. 8, the amplitude controller 708 includes a correctionamount calculator 881 and an amplitude correction unit 882. Thecorrection amount calculator 881 calculates an amplitude correctioncoefficient using a phase rotation amount transmitted by a change amountgenerator 203. The amplitude correction unit 882 multiplies adeteriorated signal amplitude spectrum supplied from a transformer 201by the calculated amplitude correction coefficient, and supplies themultiplication result to an inverse transformer 204. Multiplication ofthe amplitude correction coefficient can cancel a drop of the outputlevel when a deteriorated signal phase spectrum 220 is controlled toobtain an enhanced signal phase spectrum 240.

A drop of the output level by phase rotation in correlation eliminationwill be explained with reference to FIGS. 9 and 10.

FIGS. 9 and 10 show signals when a deteriorated signal is processed bythe arrangement shown in the block diagram of FIG. 7. The differencebetween FIGS. 9 and 10 is the presence/absence of phase rotation. FIG. 9shows signals when no phase rotation is performed. FIG. 10 shows signalswhen phase rotation is performed from frame 3.

Signals when no phase rotation is performed will be described withreference to FIG. 9. A deteriorated signal is shown at the top of FIG.9. A frame divider 301 divides the deteriorated signal into frames. Thesecond signal from the top, which is sectioned by a dotted line, is asignal after frame division. FIG. 9 shows signals of four successiveframes. The frame overlapping ratio is 50%.

A windowing unit 302 performs windowing for the signals of the dividedframes. The third signal from the top, which is sectioned by a dottedline, is a signal after windowing. In FIG. 9, weighting by a rectangularwindow is executed to clearly represent the influence of phase rotation.

After that, a Fourier transformer 303 transforms a signal into one inthe frequency domain. However, the signal in the frequency domain is notshown in FIG. 9. A lower part below the dotted line of phase rotationillustrates a signal transformed to the time domain by an inverseFourier transformer 401 of the inverse transformer 204. The fourthsignal from the top, which is sectioned by a dotted line, is a signalafter phase rotation. Note that no phase rotation is performed in FIG.9, so the signal does not change from one after windowing.

A windowing unit 402 executes windowing again for an enhanced signaloutput from the inverse Fourier transformer 401 of the inversetransformer 204. FIG. 9 shows a case in which weighting by a rectangularwindow is executed. A frame composition unit 403 composes windowedsignals. At this time, the times between frames need to be equal to eachother. Since the frame overlapping ratio is 50%, frames overlap eachother just by half. If no phase rotation is executed, the input signaland output signal coincide with each other, as shown in FIG. 9.

On the other hand, signals when the phase is rotated will be explainedwith reference to FIG. 10. FIG. 10 shows signals when phase rotation isexecuted from frame 3. The same deteriorated signal as that shown inFIG. 9 is illustrated at the top of FIG. 10. Signals after framedivision and windowing are also the same as those shown in FIG. 9.

FIG. 10 shows a case in which given phase rotation is executed fromframe 3. Attention is paid to a section indicated by a right-pointingtriangle below the dotted line of the phase rotation processing. Thephase rotation processing shifts signals of frames 3 and 4 in the timedirection. The signal having undergone phase rotation is windowed againto perform frame composition. At this time, signals of frames 2 and 3differ from each other in section ii where frames 2 and 3 overlap eachother. As a result, the output signal level after frame compositionbecomes low in section ii. That is, when phase rotation is executed, theoutput signal level drops in section ii of FIG. 10.

The drop of the output signal level caused by phase rotation can also beexplained by vector composition in the frequency domain by replacingaddition in the time domain with addition in the frequency domain.

FIG. 11 shows deteriorated signals x₁[n] and x₂[m] of two successiveframes after frame division and windowing. Assume that the overlappingratio is 50%. Note that n represents the discrete time of x₁, and mrepresents the discrete time of x₂. The overlapping ratio of 50% yields:

$\begin{matrix}{m = {n + \frac{L}{2}}} & (28)\end{matrix}$

The relationship between x₁ and x₂ is given by:

$\begin{matrix}{{x_{2}\lbrack m\rbrack} = {x_{1}\left\lbrack {n + \frac{L}{2}} \right\rbrack}} & (29)\end{matrix}$

First, a transform equation from a time domain signal into a frequencydomain signal, and an inverse transform equation will be described.Based on Fourier transform of a time domain signal x[n], a frequencydomain signal X[k] is given by:

$\begin{matrix}{{X\lbrack k\rbrack} = {\sum\limits_{n = 0}^{L - 1}{{x\lbrack n\rbrack}^{{- {j2\pi}}\frac{n}{L}k}}}} & (30)\end{matrix}$

where k represents the discrete frequency, and L represents the framelength.

Inverse transform to return the frequency domain signal X[k] into thetime domain signal x[n] is given by:

$\begin{matrix}{{x\lbrack n\rbrack} = {\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}{{X\lbrack k\rbrack}^{{j2\pi}\frac{n}{L}k}}}}} & (31)\end{matrix}$

Based on this, transform of the time domain signals x₁[n] and x₂[m] intofrequency domain signals X₁[k] and X₂[k], respectively, is given by:

$\begin{matrix}{{X_{1}\lbrack k\rbrack} = {\sum\limits_{n = 0}^{L - 1}{{x_{1}\lbrack n\rbrack}^{{- {j2\pi}}\frac{n}{L}k}}}} & (32) \\{{X_{2}\lbrack k\rbrack} = {\sum\limits_{m = 0}^{L - 1}{{x_{2}\lbrack m\rbrack}^{{- {j2\pi}}\frac{m}{L}k}}}} & (33)\end{matrix}$

Based on equation (31), inverse transform to return the frequency domainsignals X₁[k] and X₂[k] into the time domain signals x₁[n] and x₂[m],respectively, is given by:

$\begin{matrix}{{x_{1}\lbrack n\rbrack} = {\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}{{X_{1}\lbrack k\rbrack}^{{j2\pi}\frac{n}{L}k}}}}} & (34) \\{{x_{2}\lbrack m\rbrack} = {\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}{{X_{2}\lbrack k\rbrack}^{{j2\pi}\frac{m}{L}k}}}}} & (35)\end{matrix}$

The inverse Fourier transformer 401 transforms a frequency domain signalinto a time domain signal by inverse Fourier transform. After that, theframe composition unit 403 performs overlap addition of enhanced soundsof preceding and current frames.

For example, at the overlapping ratio of 50% in the illustrated example,the frame composition unit 403 adds adjacent frames in a section of thediscrete time m=L/2 to L−1. The addition section m=L/2 to L−1 will beconsidered.

Substituting equations (34) and (35) into time domain addition yields:

$\begin{matrix}{{{x_{1}\lbrack n\rbrack} + {x_{2}\lbrack m\rbrack}} = {{\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}{{X_{1}\lbrack k\rbrack}^{{j2\pi}\frac{n}{L}k}}}} + {\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}{{X_{2}\lbrack k\rbrack}^{{j2\pi}\frac{m}{L}k}}}}}} & (36)\end{matrix}$

Furthermore, substituting equations (32) and (33) into the frequencydomain signals X₁[k] and X₂[k] in equation (36) yields:

$\begin{matrix}\begin{matrix}{{{x_{1}\lbrack n\rbrack} + {x_{2}\lbrack m\rbrack}} = {{\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}{{X_{1}\lbrack k\rbrack}^{{j2\pi}\; \frac{n}{L}k}}}} + {\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}{{X_{2}\lbrack k\rbrack}^{{j2\pi}\; \frac{m}{L}k}}}}}} \\{= {{\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}{\left( {\sum\limits_{n = 0}^{L - 1}{{x_{1}\lbrack n\rbrack}^{{- {j2\pi}}\; \frac{n}{L}k}}} \right)^{{j2\pi}\; \frac{n}{L}k}}}} +}} \\{{\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}{\left( {\sum\limits_{m = 0}^{L - 1}{{x_{2}\lbrack m\rbrack}^{{- {j2\pi}}\; \frac{m}{L}k}}} \right)^{{j2\pi}\frac{m}{L}k}}}}}\end{matrix} & (37)\end{matrix}$

Equation (37) is expanded into:

$\begin{matrix}{{{x_{1}\lbrack n\rbrack} + {x_{2}\lbrack m\rbrack}} = {{{\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}{\left( {\sum\limits_{n = 0}^{L - 1}{{x_{1}\lbrack n\rbrack}^{{- {j2\pi}}\; \frac{n}{L}k}}} \right)^{{j2\pi}\; \frac{n}{L}k}}}} + {\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}{\left( {\sum\limits_{m = 0}^{L - 1}{{x_{2}\lbrack m\rbrack}^{{- {j2\pi}}\; \frac{m}{L}k}}} \right)^{{j2\pi}\; \frac{m}{L}k}}}}} = {{{\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}{\left( {{{x_{1}\lbrack 0\rbrack}^{{- {j2\pi}}\; \frac{0}{L}k}} + {{x_{1}\lbrack 1\rbrack}^{{- {j2\pi}}\; \frac{1}{L}k}} + \mspace{11mu} \ldots \mspace{11mu} + {{x_{1}\left\lbrack {L - 1} \right\rbrack}^{{- {j2\pi}}\; \frac{L - 1}{L}k}}} \right)^{{j2\pi}\; \frac{n}{L}k}}}} + {\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}{\left( {{{x_{2}\lbrack 0\rbrack}^{{- {j2\pi}}\; \frac{0}{L}k}} + {{x_{2}\lbrack 1\rbrack}^{{- {j2\pi}}\; \frac{1}{L}k}} + \mspace{11mu} \ldots \mspace{11mu} + {{x_{2}\left\lbrack {L - 1} \right\rbrack}^{{- {j2\pi}}\; \frac{L - 1}{L}k}}} \right)^{{j2\pi}\; \frac{m}{L}k}}}}} = {{\frac{1}{L}\left\{ {{{x_{1}\lbrack 0\rbrack}{\sum\limits_{k = 0}^{L - 1}^{j\frac{2\pi}{L}\; {({n - 0})}k}}} + {{x_{1}\lbrack 1\rbrack}{\sum\limits_{k = 0}^{L - 1}^{j\frac{2\pi}{L}{({n - 1})}\; k}}} + \mspace{11mu} \ldots \mspace{11mu} + {{x_{1}\left\lbrack {L - 1} \right\rbrack}{\sum\limits_{k = 0}^{L - 1}^{j\frac{2\pi}{L}{({n - L + 1})}k}}}} \right\}} + {\frac{1}{L}\left\{ {{{x_{2}\lbrack 0\rbrack}{\sum\limits_{k = 0}^{L - 1}^{j\frac{2\pi}{L}\; {({m - 0})}k}}} + {{x_{2}\lbrack 1\rbrack}{\sum\limits_{k = 0}^{L - 1}^{j\frac{2\pi}{L}{({m - 1})}\; k}}} + \mspace{11mu} \ldots \mspace{11mu} + {{x_{2}\left\lbrack {L - 1} \right\rbrack}{\sum\limits_{k = 0}^{L - 1}^{j\frac{2\pi}{L}{({m - L + 1})}k}}}} \right\}}}}}} & (38)\end{matrix}$

Total sum calculation included in each term of equation (38) will beconsidered. Introducing an arbitrary integer g establishes:

$\begin{matrix}{\sum\limits_{k = 0}^{L - 1}^{j\frac{2\pi}{L}{gk}}} & (39)\end{matrix}$

An inverse Fourier transform of a delta function δ[g] is given by:

$\begin{matrix}{{\delta \lbrack g\rbrack} = {\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}^{j\frac{2\pi}{L}{gk}}}}} & (40)\end{matrix}$

The delta function δ[g] is given by:

$\begin{matrix}{{\delta \lbrack g\rbrack} = \left\{ \begin{matrix}1 & {g = 0} \\0 & {g \neq 0}\end{matrix} \right.} & (41)\end{matrix}$

Based on equation (40), equation (39) can be rewritten into:

$\begin{matrix}{{\sum\limits_{k = 0}^{L - 1}^{j\frac{2\pi}{L}{gk}}} = {L \cdot {\delta \lbrack g\rbrack}}} & (42)\end{matrix}$

From the relation of equation (42), equation (38) can be rewritten into:

$\begin{matrix}{{{x_{1}\lbrack n\rbrack} + {x_{2}\lbrack m\rbrack}} = {{\frac{1}{L}\left\{ {{{L \cdot {x_{1}\lbrack 0\rbrack}}{\delta \lbrack 0\rbrack}} + {{L \cdot {x_{1}\lbrack 1\rbrack}}{\delta \left\lbrack {n - 1} \right\rbrack}} + \; \ldots \; + {{L \cdot {x_{1}\left\lbrack {L - 1} \right\rbrack}}{\delta \left\lbrack {n - L + 1} \right\rbrack}}} \right\}} + {\frac{1}{L}\left\{ {{{L \cdot {x_{2}\lbrack 0\rbrack}}{\delta \lbrack 0\rbrack}} + {{L \cdot {x_{2}\lbrack 1\rbrack}}{\delta \left\lbrack {m - 1} \right\rbrack}} + \; \ldots \; + {{L \cdot {x_{2}\left\lbrack {L - 1} \right\rbrack}}{\delta \left\lbrack {m - L + 1} \right\rbrack}}} \right\}}}} & (43)\end{matrix}$

Hence, equation (38) is rewritten into:

$\begin{matrix}\begin{matrix}{{{x_{1}\lbrack n\rbrack} + {x_{2}\lbrack m\rbrack}} = {{\frac{1}{L}\left\{ {L \cdot {x_{1}\lbrack n\rbrack}} \right\}} + {\frac{1}{L}\left\{ {L \cdot {x_{2}\lbrack m\rbrack}} \right\}}}} \\{= {{x_{1}\lbrack n\rbrack} + {x_{2}\lbrack m\rbrack}}}\end{matrix} & (44)\end{matrix}$

A case in which phase rotation is performed for the frequency domainsignal X₂[k] will be considered. A time domain signal at this time is asshown in FIG. 12.

When the phase spectrum of X₂[k] is rotated by φ[k], inverse transformis given by:

$\begin{matrix}{{x_{2}\lbrack m\rbrack} = {\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}\; {{X_{2}\lbrack k\rbrack}^{{j\varphi}{\lbrack k\rbrack}}^{{j2\pi}\frac{m}{L}k}}}}} & (45)\end{matrix}$

Substituting equation (45) into equation (36) yields:

$\begin{matrix}\begin{matrix}{{{x_{1}\lbrack n\rbrack} + {x_{2}\lbrack m\rbrack}} = {{\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}\; {{X_{1}\lbrack k\rbrack}^{{j2\pi}\frac{n}{L}k}}}} + {\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}\; {{X_{2}\lbrack k\rbrack}^{{j\varphi}{\lbrack k\rbrack}}^{{j2\pi}\frac{m}{L}k}}}}}} \\{= {{\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}\; {\left( {\sum\limits_{n = 0}^{L - 1}\; {{x_{1}\lbrack n\rbrack}^{{- {j2\pi}}\frac{n}{L}k}}} \right)^{{j2\pi}\frac{n}{L}k}}}} +}} \\{{\frac{1}{L}{\sum\limits_{k = 0}^{L - 1}\; {\left( {\sum\limits_{m = 0}^{L - 1}\; {{x_{2}\lbrack m\rbrack}^{- {({{{j2\pi}\frac{m}{L}k} + {\varphi {\lbrack k\rbrack}}})}}}} \right)^{{j2\pi}\frac{m}{L}k}}}}}\end{matrix} & (46)\end{matrix}$

Equation (46) is expanded into:

$\begin{matrix}\begin{matrix}{{{x_{1}\lbrack n\rbrack} + {x_{2}\lbrack m\rbrack}} = {\frac{1}{L}\left\{ {{{x_{1}\lbrack 0\rbrack}{\sum\limits_{k = 0}^{L - 1}\; ^{j\frac{2\pi}{L}{({n - 0})}k}}} + {{x_{1}\lbrack 1\rbrack}{\sum\limits_{k = 0}^{L - 1}\; ^{j\frac{2\pi}{L}{({n - 1})}k}}} + \ldots +} \right.}} \\\left. {{x_{1}\left\lbrack {L - 1} \right\rbrack}{\sum\limits_{k = 0}^{L - 1}\; ^{j\frac{2\; n}{L}{({n - L + 1})}k}}} \right\} \\{= {\frac{1}{L}\left\{ {{{x_{2}\lbrack 0\rbrack}{\sum\limits_{k = 0}^{L - 1}\; {^{j\frac{2\pi}{L}{({m - 0})}k}^{{j\varphi}{\lbrack k\rbrack}}}}} +} \right.}} \\{{{{x_{2}\lbrack 1\rbrack}{\sum\limits_{k = 0}^{L - 1}\; {^{j\frac{2\pi}{L}{({m - 1})}k}^{{j\varphi}{\lbrack k\rbrack}}}}} + \ldots +}} \\\left. {{x_{2}\left\lbrack {L - 1} \right\rbrack}{\sum\limits_{k = 0}^{L - 1}\; {^{j\frac{2\pi}{L}{({m - L + 1})}k}^{{j\varphi}{\lbrack k\rbrack}}}}} \right\}\end{matrix} & (47)\end{matrix}$

Assuming that the overlapping ratio is 50%, the overlapping sectionn=L/2 to L−1 will be considered. In the overlapping section, equation(47) can be expanded into:

$\begin{matrix}\begin{matrix}{{{x_{1}\left\lbrack {n + \frac{L}{2}} \right\rbrack} + {x_{2}\lbrack m\rbrack}} = {\frac{1}{L}\left\{ {{{x_{1}\left\lbrack \frac{L}{2} \right\rbrack}{\sum\limits_{k = 0}^{L - 1}\; ^{j\frac{2\pi}{L}{({m + \frac{L}{2} - \frac{L}{2}})}k}}} +} \right.}} \\{{{{x_{1}\left\lbrack {\frac{L}{2} + 1} \right\rbrack}{\sum\limits_{k = 0}^{L - 1}\; ^{j\frac{2\pi}{L}{({m - \frac{L}{2} - {1\frac{L}{2}} + 1})}k}}} + \ldots +}} \\{\left. {{x_{1}\left\lbrack {L - 1} \right\rbrack}{\sum\limits_{k = 0}^{L - 1}\; ^{j\frac{2\pi}{L}{({m + \frac{L}{2} - L + 1 - L - 1})}k}}} \right\} +} \\{{\frac{1}{L}\left\{ {{{x_{2}\lbrack 0\rbrack}{\sum\limits_{k = 0}^{L - 1}\; {^{j\frac{2\pi}{L}{({n - 0})}k}^{{j\varphi}{\lbrack k\rbrack}}}}} +} \right.}} \\{{{{x_{1}\lbrack 1\rbrack}{\sum\limits_{k = 0}^{L - 1}\; {^{j\frac{2\pi}{L}{({n - 1})}k}^{{j\varphi}{\lbrack k\rbrack}}}}} + \ldots +}} \\\left. {{x_{2}\left\lbrack {L - \frac{L}{2} - 1} \right\rbrack}{\sum\limits_{k = 0}^{L - 1}\; {^{j\frac{2\pi}{L}{({n + \frac{L}{2} - L + 2})}k}^{{j\varphi}{\lbrack k\rbrack}}}}} \right\} \\{= {\frac{1}{L}\left\{ {{{x_{2}\lbrack 0\rbrack}{\sum\limits_{k = 0}^{L - 1}\; ^{j\frac{2\pi}{L}{nk}}}} + {{x_{2}\lbrack 1\rbrack}{\sum\limits_{k = 0}^{L - 1}\; ^{j\frac{2\pi}{L}{nk}}}} + \ldots +} \right.}} \\{\left. {{x_{1}\left\lbrack {L - 1} \right\rbrack}{\sum\limits_{k = 0}^{L - 1}\; ^{j\frac{2\pi}{L}{nk}}}} \right\} +} \\{{\frac{1}{L}\left\{ {{{x_{2}\lbrack 0\rbrack}{\sum\limits_{k = 0}^{L - 1}\; {^{j\frac{2\pi}{L}{({n - 0})}k}^{{j\varphi}{\lbrack k\rbrack}}}}} +} \right.}} \\{{{{x_{2}\lbrack 1\rbrack}{\sum\limits_{k = 0}^{L - 1}\; {^{j\frac{2\pi}{L}{({n - 1})}k}^{{j\varphi}{\lbrack k\rbrack}}}}} + \ldots +}} \\\left. {{x_{2}\left\lbrack {L - \frac{L}{2} - 1} \right\rbrack}{\sum\limits_{k = 0}^{L - 1}\; {^{j\frac{2\pi}{L}{({n - \frac{L}{2} - L + 3})}k}^{{j\varphi}{\lbrack k\rbrack}}}}} \right\} \\{= {\frac{1}{L}\left\{ {{{x_{3}\lbrack 0\rbrack}{\sum\limits_{k = 0}^{L - 1}\; {^{j\frac{2\pi}{L}{nk}}\left( {1 + ^{{j\varphi}{\lbrack k\rbrack}}} \right)}}} +} \right.}} \\{{{{x_{2}\lbrack 1\rbrack}{\sum\limits_{k = 0}^{L - 1}\; {^{j\frac{2\pi}{L}{({n - 1})}k}\left( {1 + ^{{j\varphi}{\lbrack k\rbrack}}} \right)}}} + \ldots +}} \\{{{x_{2}\left\lbrack {\frac{L}{2} - 1} \right\rbrack}{\sum\limits_{k = 0}^{L - 1}\; {^{j\frac{2\pi}{L}{({n - \frac{L}{2} - 1})}k}\left( {1 + ^{{j\varphi}{\lbrack k\rbrack}}} \right)}}}}\end{matrix} & (48)\end{matrix}$

In this case, the parenthesized term (given by expression (49) below) ineach term is vector composition.

1+e ^(jφ[k])  (49)

When attention is paid to a specific frequency k, a frequency domainsignal can be represented as shown in FIG. 13.

When no phase rotation is performed, that is, φ[k]=0, a frequency domainsignal is as shown in FIG. 14.

The absolute value of expression (49) is given by:

$\begin{matrix}\begin{matrix}{{{1 + ^{{j\varphi}{\lbrack k\rbrack}}}} = {{1 + {\cos \; {\varphi \lbrack k\rbrack}} + {j\; \sin \; {\varphi \lbrack k\rbrack}}}}} \\{= \sqrt{\left( {1 + {\cos \; {\varphi \lbrack k\rbrack}}} \right)^{2} + {\sin^{2}{\varphi \lbrack k\rbrack}}}} \\{= \sqrt{1 + {2\; \cos \; {\varphi \lbrack k\rbrack}} + {\cos^{2}{\varphi \lbrack k\rbrack}} + {\sin^{2}{\varphi \lbrack k\rbrack}}}} \\{= \sqrt{2\left( {1 + {\cos \; {\varphi \lbrack k\rbrack}}} \right)}}\end{matrix} & (50)\end{matrix}$

A condition under which the absolute value indicated by expression (49)is maximized is φ[k]=0, and the absolute value is 2. That is, executingphase rotation decreases the magnitude of an output signal.

To correct the drop of the output signal level, the correction amountcalculator 881 decides the amplitude correction amount of the enhancedsignal amplitude spectrum.

A correction amount calculation method will be explained in detail. Tosimplify a problem, attention is paid to variations of the magnitudecaused by phase rotation, and respective frequency components areassumed to have been normalized to unit vectors.

First, a case in which no phase rotation is performed will beconsidered. A composite vector when the phase is the same betweensuccessive frames is represented by a vector S shown in FIG. 14. Themagnitude |S| of this vector is given by:

$\begin{matrix}\begin{matrix}{{S} = \sqrt{\left\{ {1 + 1} \right\}^{2}}} \\{= \sqrt{2^{2}}} \\{= 2}\end{matrix} & (51)\end{matrix}$

On the other hand, if phase rotation is performed by a normal randomnumber, a composite vector when the phase difference between successiveframes is φ is represented by a vector S′ shown in FIG. 13. Themagnitude |S′| of this vector is given by:

|S′|=√{square root over ({1+cos φ}²+{sin φ}²)}=√{square root over(2+2{cos φ})}  (52)

An expected value E(|S′|²) is given by:

E(|S′| ²)=E(2+2 cos φ)=E(2)+(2 cos φ)  (53)

For a normal random number, the rate of occurrence of φ is decided by anormal distribution. To obtain an expected power value when phaserotation is performed by a normal random number, therefore, it isnecessary to perform weighting based on the rate of occurrence of φ.

More specifically, a weighting function f(φ) based on the rate ofoccurrence of φ is introduced. The weighting function f(φ) weightscos(φ). Furthermore, normalization using the integral value of theweighting function f(φ) can provide an expected power value.

By introducing the weighting function f(φ) and its integral value intoequation (53) which expresses an expected output power value based on auniform normal random number, an expected output power value E(S′²) whenphase rotation is performed using a normal random number is given by:

$\begin{matrix}{{E\left( {S^{\prime 2}} \right)} = {{E(2)} + {E\left( {2\frac{f(\varphi)}{\int_{- \pi}^{\pi}{{f(\varphi)}\ {\varphi}}}{\cos (\varphi)}} \right)}}} & (54)\end{matrix}$

Expressing the weighting function f(φ) by a normal distribution yields:

$\begin{matrix}{{f(\varphi)} = {\frac{1}{\sqrt{2\pi}\sigma}{\exp \left( {- \frac{\left( {\varphi - \mu} \right)^{2}}{2\sigma^{2}}} \right)}}} & (55)\end{matrix}$

where σ represents the variance, and μ represents the average.

For example, for a standardized normal distribution having the averageμ=0 and the variance σ=1, the weighting function f(φ) is given by:

$\begin{matrix}{{f(\varphi)} = {\frac{1}{\sqrt{2\pi}}{\exp \left( {- \frac{\varphi^{2}}{2}} \right)}}} & (56)\end{matrix}$

Substituting equation (56) into equation (54) yields:

$\begin{matrix}{{E\left( {S^{\prime 2}} \right)} = {{E(2)} + {E\left( {2\frac{\exp \left( {- \frac{\varphi^{2}}{2}} \right)}{\int_{- \pi}^{\pi}{{\exp \left( {- \frac{\varphi^{2}}{2}} \right)}\ {\varphi}}}{\cos (\varphi)}} \right)}}} & (57)\end{matrix}$

Then, numerically calculating the second term on the right-hand side ofequation (57) establishes:

E(|S′| ²)=2{1+0.609}=3.219  (58)

The ratio of E(|S′|²) to E(|S²|) when no phase rotation is performed isgiven by:

E(|S′| ²)/E(|S′| ²)=3.218/4=0.805  (59)

When rotating the phase with a normal random number of the standardizednormal distribution, the correction amount calculator 881 setssqrt(1/0.805) as the correction coefficient, and transmits it to theamplitude correction unit 882. The phase controller 202 may performphase rotation for all or some frequencies. The amplitude controller 708performs amplitude correction for only frequencies having undergonephase rotation. Therefore, a correction coefficient for frequencies notto undergo phase rotation is set to 1.0. Only a correction coefficientfor frequencies having undergone phase rotation is a value derived here.

Although not all the phase rotation characteristics can be completelyexpressed by a normal distribution, it is possible to apply theabove-described correction amount calculation method by approximationusing the normal distribution. To do this, it is necessary to collectstatistics based on the value of a change amount generated by the changeamount generator 203 and its appearance frequency, and obtain theaverage μ and variance σ of the normal distribution indicated byequation (55). Subsequently, calculation from equation (55) to equation(59) is performed to obtain the square root of the reciprocal of thecalculation result as a correction coefficient.

As described above, the noise suppression apparatus 700 according tothis embodiment can cause the amplitude controller 708 to eliminate theinfluence of control of the phase spectrum on an output signal level.Thus, the noise suppression apparatus 700 can obtain a high-qualityenhanced signal.

Fifth Embodiment

A noise suppression apparatus 1500 according to the fifth embodiment ofthe present invention will be described with reference to FIG. 15. Inthis embodiment, the noise suppression apparatus 1500 is different fromthat in the third embodiment in that an amplitude controller 708 isincluded. The components except for the amplitude controller 708 are thesame as in the third embodiment, and the amplitude controller 708 is thesame as in the fourth embodiment. The same reference numerals denote thesame components and a detailed description thereof will be omitted.

According to this embodiment, it is possible to efficiently suppressnoise derived from a phase by rotating or replacing a deterioratedsignal phase spectrum using a deteriorated signal amplitude spectrum ora value obtained from it, and suppress a decrease in output level causedby phase control by controlling an amplitude.

Sixth Embodiment

A noise suppression apparatus 1600 according to the sixth embodiment ofthe present invention will be described with reference to FIG. 16. Thisembodiment is different from the second embodiment in that the upperlimit of a phase rotation amount is limited. The remaining componentsand operations are the same as those in the second embodiment and adetailed description thereof will be omitted.

FIG. 16 is a block diagram showing the arrangement of the noisesuppression apparatus 1600 according to this embodiment. As shown inFIG. 16, the noise suppression apparatus 1600 according to thisembodiment includes a change amount limiter 1601 in addition to a changeamount generator 203 and phase controller 202 which have been describedin the second embodiment. The change amount generator 203 generates achange amount of a deteriorated signal phase spectrum, and supplies itto the phase controller 202 while being limited by the change amountlimiter 1601.

The change amount limiter 1601 limits the rotation amount generated bythe change amount generator 203 to a given range. That is, the changeamount limiter 1601 limits the distribution of φ to an arbitrary rangefrom 0 to 2π. For example, the change amount limiter 1601 limits thedistribution of φ to a range from 0 to π/2. This causes the features ofthe deteriorated signal phase spectrum to remain in the enhanced signalphase spectrum to some extent. The features of the deteriorated signalare held to some extent, as compared with a case in which the phase iscompletely rotated at random, thereby reducing the influence on a targetsound. This reduces the distortion of the target sound.

According to this embodiment of the present invention, in addition tothe effects of the second embodiment, deterioration of a target soundcan be reduced by limiting a phase rotation amount.

Seventh Embodiment

The seventh embodiment of the present invention will be described withreference to FIG. 17. This embodiment according to the present inventionis different from the fourth embodiment in that a phase component isdelayed to obtain a phase component difference between frames, and thena correction amount is calculated from the difference. That is, thisembodiment is different from the second embodiment in the internalarrangement of an amplitude controller 1708. The remaining componentsand operations are the same as those in the second embodiment and adescription thereof will be omitted.

FIG. 17 is a block diagram showing the arrangement of the amplitudecontroller 1708 according to this embodiment. As shown in FIG. 17, aphase controller 202 according to this embodiment supplies a phase afterrotation to the amplitude controller 1708. The amplitude controller 1708includes a phase component delay unit 1781, a correction amountcalculator 1782, and an amplitude correction unit 882.

The phase component delay unit 1781 holds an enhanced signal phasespectrum supplied from the phase controller 202 for one frame, andsupplies it to the correction amount calculator 1782.

The correction amount calculator 1782 calculates an amplitude correctionamount based on the enhanced signal phase spectrum of an immediatelypreceding frame supplied from the phase component delay unit 1781 andthe current enhanced signal phase spectrum supplied from the phasecontroller 202, and then transmits the calculated amplitude correctionamount to the amplitude correction unit 882.

According to this embodiment, in addition to the effects of the secondembodiment, it is possible to correct an output level even if theexpected value of the output level cannot be derived mathematicallybased on a phase change amount.

The correction amount calculator 1782 obtains the magnitude of acomposite vector at each frequency from the enhanced signal phasespectra of preceding and current frames, and decides a correctioncoefficient based on the magnitude. Letting α be the phase of apreceding frame and be the phase of a current frame, a magnitude |S′| ofa composite vector is given by:

|S′|=√{square root over ({cos α+cos β}²+{sin α+sin β}²)}=√{square rootover (2+2{sin α sin β}+2{cos α cos β})}  (60)

A magnitude |S| of a composite vector when the phases of successiveframes coincide with each other is |S|=2, which has already been derivedby equation (51). Therefore, the amplitude correction amount is givenby:

|S|/|S′|=2/√{square root over (2+2{sin α sin β}+2{cos α cos β})}  (61)

In this embodiment, this value is supplied to the amplitude controller1708 to correct the enhanced signal amplitude spectrum, therebycanceling a drop of the output level. In this embodiment, the componentsand operations except for the phase rotation unit are the same as thosein the second embodiment, and a description thereof will be omitted.

Eighth Embodiment

The eighth embodiment of the present invention will be explained withreference to FIG. 18. FIG. 18 is a block diagram showing thearrangements of a phase controller 202 and amplitude controller 1808according to this embodiment.

An invention according to this embodiment is different from FIG. 8 (thefourth embodiment) in that an input/output ratio calculator 1881 isincluded. The input/output ratio calculator 1881 receives a deterioratedsignal from an input terminal 206 and an enhanced signal from an inversetransformer 204, and calculates an input/output level ratio. Theinput/output ratio calculator 1881 supplies the input/output level ratioto a correction amount calculator 1882. The correction amount calculator1882 calculates a correction amount so that the level of the enhancedsignal becomes equal to that of the deteriorated signal. An amplitudecorrection unit 882 corrects an enhanced signal amplitude spectrum bythe calculated correction amount.

The input/output ratio calculator 1881 obtains the level ratio from thedeteriorated signal and the time domain signal of the enhanced signal.

A level ratio R of an enhanced signal x_(n)(t) of the nth frame to adeteriorated signal y_(n)(t) of the nth frame is given by:

$\begin{matrix}{R = {\sum\limits_{t = 0}^{L - 1}\; {{x_{n}(t)}/{\sum\limits_{t = 0}^{L - 1}\; {y_{n}(t)}}}}} & (62)\end{matrix}$

where t represents the sample time, and L represents the frame length ofFourier transform.

The correction amount calculator 1882 obtains an amplitude correctionamount G from the ratio value R and the number of frequency componentshaving undergone phase rotation. When a transformer divides the timedomain signal into N frequency components and phase rotation isperformed for M phase spectra, the amplitude correction amount G isgiven by:

$\begin{matrix}{G = \frac{M}{{N\left( {R - 1} \right)} + M}} & (63)\end{matrix}$

The amplitude correction unit 882 executes amplitude correction for onlyfrequencies having undergone phase rotation based on phase rotationpresence/absence information transmitted from a change amount generator203. In this embodiment, the components and operations except for theinput/output ratio calculator 1881 and correction amount calculator 1882are the same as those in the fourth embodiment, and a descriptionthereof will be omitted.

According to this embodiment of the present invention, a correctioncoefficient is obtained from a time domain signal, so the output levelcan be corrected regardless of a phase replacement amount decisionmethod.

Ninth Embodiment

The ninth embodiment of the present invention will be described withreference to FIG. 19. FIG. 19 is a block diagram showing the arrangementof an amplitude controller 1908 according to this embodiment. As shownin FIG. 19, the amplitude controller 1908 according to this embodimentincludes an averaging processor 1981, in addition to an input/outputratio calculator 1881 included in the eighth embodiment. The componentsand operations except for the averaging processor 1981 are the same asthose in the eighth embodiment, and a description thereof will beomitted.

The averaging processor 1981 receives a deteriorated signal from aninput terminal 206, performs averaging processing, and then supplies anaverage value to the input/output ratio calculator 1881. The averagingprocessor 1981 receives an enhanced signal from an inverse transformer204, performs averaging processing, and then supplies an average valueto the input/output ratio calculator 1881. The input/output ratiocalculator 1881 receives the average values of the deteriorated signaland enhanced signal from the averaging processor 1981, and calculatesthe level ratio.

The averaging processor 1981 averages the levels of the deterioratedsignal and enhanced signal for an arbitrary time length. Morespecifically, the averaging processor 1981 averages the levels of thedeteriorated signal and enhanced signal using a moving average, leakageintegral, or the like.

According to this embodiment of the present invention, in addition tothe effects of the eighth embodiment, since an averaged level is used,variations of the correction amount can be suppressed to improve thequality of an output signal.

10th Embodiment

The 10th embodiment of the present invention will be described withreference to FIGS. 20 and 21. FIG. 20 is a block diagram showing thearrangement of a noise suppression apparatus 2000 according to thisembodiment. The noise suppression apparatus 2000 according to thisembodiment includes an amplitude component delay unit 2011, a phasecomponent delay unit 2012, and an inverse transformer 2013, in additionto the arrangement shown in FIG. 2 of the second embodiment. Thisembodiment is also different from the second embodiment in the internalarrangement of an amplitude controller 2008. In this embodiment,operations except for those of the amplitude component delay unit 2011,phase component delay unit 2012, and amplitude controller 2008 are thesame as those in the second embodiment, and a description thereof willbe omitted.

A deteriorated signal 210 supplied to an input terminal 206 is suppliedto a transformer 201 and the amplitude controller 2008. The transformer201 supplies a deteriorated signal amplitude spectrum 230 to theamplitude component delay unit 2011 and inverse transformer 2013. Thetransformer 201 also supplies a deteriorated signal phase spectrum 220to a phase controller 202 and a change amount generator 203. The phasecontroller 202 controls the deteriorated signal phase spectrum 220supplied from the transformer 201 by using a change amount generated bythe change amount generator 203, and supplies it as an enhanced signalphase spectrum to the inverse transformer 2013 and phase component delayunit 2012. Also, the change amount generator 203 transmits thepresence/absence of phase rotation at each frequency to the amplitudecontroller 2008.

By using the deteriorated signal amplitude spectrum 230 supplied fromthe transformer 201 and the deteriorated signal phase spectrum suppliedfrom the phase controller 202, the inverse transformer 2013 transmits,to the amplitude controller 2008, a signal whose level has dropped dueto phase rotation.

The amplitude component delay unit 2011 delays the deteriorated signalamplitude spectrum 230 supplied from the transformer 201, and suppliesit to the amplitude controller 2008.

The phase component delay unit 2012 delays the enhanced signal phasespectrum supplied from the phase controller 202, and supplies it to aninverse transformer 204. The amplitude controller 2008 generates acorrected amplitude spectrum 250 based on the deteriorated signalamplitude spectrum supplied from the amplitude component delay unit 2011by using the output of the inverse transformer 2013 and the deterioratedsignal 210.

The inverse transformer 204 performs inverse transform by composing anenhanced signal phase spectrum 240 supplied from the phase controller202 via the phase component delay unit 2012 and the corrected amplitudespectrum 250 supplied from the amplitude controller 2008, and suppliesthe inverse transform result as an enhanced signal to an output terminal207.

The phase controller 202 controls the deteriorated signal phase spectrum220, and the inverse transformer 2013 transforms the deteriorated signalphase spectrum 220 into a time domain signal. By using this signal andthe deteriorated signal 210, the amplitude controller 2008 obtains theamount of a variation in level caused by phase rotation.

Since the variation amount arises from a variation caused by onlyrotation processing by the phase controller 202. The amplitudecontroller 2008 can, therefore, accurately grasp a variation in levelcaused by phase rotation. Although the amplitude controller 2008executes amplitude correction using the level ratio, the obtained levelratio is one for an immediately preceding frame. Thus, the amplitudecomponent delay unit 2011 and phase component delay unit 2012 areintroduced, and the amplitude controller 2008 performs amplitudecorrection for the frequency component of an immediately precedingframe.

FIG. 21 is a block diagram for explaining the internal arrangements ofthe phase controller 202 and amplitude controller 2008 according to thisembodiment. An input/output ratio calculator 2181 calculates a levelratio from a deteriorated signal supplied from the input terminal 206and a signal which is supplied from the inverse transformer 2013 andcontains a level drop caused by phase rotation, and supplies the levelratio to a correction amount calculator 2182.

The correction amount calculator 2182 receives phase rotationpresence/absence information at each frequency from the change amountgenerator 203, and calculates an amplitude correction amount. Anamplitude correction unit 882 corrects the enhanced signal phasespectrum at each frequency based on the amplitude correction amount, andsupplies it to the inverse transformer 204.

In addition to the effects of the eighth embodiment, the noisesuppression apparatus 2000 according to this embodiment can avoid adelay of the input/output ratio that is inevitable in the eighth andninth embodiments, thereby correcting the output level more accurately.

11th Embodiment

The 11th embodiment of the present invention will be described withreference to FIG. 22. As shown in FIG. 22, a noise suppression apparatus2200 according to this embodiment includes a frame overlappingcontroller 2208, in addition to the arrangement of the fourthembodiment. The frame overlapping controller 2208 controls anoverlapping ratio when frame division and composition are performed in atransformer 201 and inverse transformer 204. The frame overlappingcontroller 2208 supplies the overlapping ratio to an amplitudecontroller 708. As described above, overlapping causes a level drop inphase rotation. The amount of level drop changes depending on theoverlapping ratio. As the overlapping ratio increases, the drop amountalso increases. Therefore, when the overlapping ratio changes, it isnecessary to control an amplitude correction amount. More specifically,the correction amount is obtained by using, as a reference, an amplitudecorrection amount G for an overlapping ratio of 50%.

When the overlapping ratio is 0%, no amplitude correction is necessary.When the overlapping ratio is 50%, the amplitude correction amount is G.Thus, by using the ratio between a frame length L and an overlappinglength Q, the amplitude correction amount is given by:

$\begin{matrix}{G^{\prime} = {{{\left( {1 - \frac{2\; Q}{L}} \right) \cdot 1} + {\frac{2\; Q}{L}G}} = {1 + {\frac{2\; Q}{L}\left( {G - 1} \right)}}}} & (64)\end{matrix}$

where G′ represents the amplitude correction amount when correctionbased on the overlapping ratio is performed.

For example, the fact that Q=L/2 holds for the overlapping ratio of 50%yields:

$\begin{matrix}{G^{\prime} = {{1 + {\frac{2\frac{L}{2}}{L}\left( {G - 1} \right)}} = {{1 + G - 1} = G}}} & (65)\end{matrix}$

The fact that Q=L/4 holds for the overlapping ratio of 25% yields:

$\begin{matrix}{G^{\prime} = {{1 + {\frac{2\frac{L}{4}}{L}\left( {G - 1} \right)}} = {{1 + {\frac{1}{2}G} - \frac{1}{2}} = {\frac{1}{2} + {\frac{1}{2}G}}}}} & (66)\end{matrix}$

The amplitude controller 708 corrects a correction coefficienttransmitted from a phase controller 202 based on equation (64), andcorrects an enhanced signal amplitude spectrum. In this embodiment, thecomponents and operations except for the frame overlapping controller2208 are the same as those in the fourth embodiment, and a descriptionthereof will be omitted.

In addition to the effects of the fourth embodiment, the noisesuppression apparatus 2200 according to this embodiment can freely setthe frame overlapping ratio.

OTHER EMBODIMENTS

Although the aforementioned first to 11th embodiments have described thenoise suppression apparatuses having different features, a noisesuppression apparatus having a combination of these features is alsoincorporated in the scope of the present invention.

The present invention can be applied to a system including pluraldevices or a single apparatus. The present invention can be applied to acase in which a software signal processing program for implementing thefunctions of the embodiments is supplied to the system or apparatusdirectly or from a remote site. Therefore, the program installed in acomputer to implement the functions of the present invention by thecomputer, a medium storing the program, or a WWW server to download theprogram is also incorporated in the scope of the present invention.

FIG. 23 is a block diagram showing the arrangement of a computer 2300which executes a signal processing program when the first embodiment isimplemented by the signal processing program. The computer 2300 includesan input unit 2301, a CPU 2302, an output unit 2303, and a memory 2304.

The CPU 2302 controls the operation of the computer 2300 by loading thesignal processing program. That is, the CPU 2302 executes the signalprocessing program stored in the memory 2304, and transforms a mixedsignal, in which the first and second signals coexist, into a phasecomponent for each frequency and an amplitude component or powercomponent for each frequency (step S2311). Then, the CPU 2302 generatesa change amount of a phase component at a predetermined frequency byusing a series of data with a cross-correlation weaker than that of thephase components and randomness lower than that of random numbers (stepS2312). The CPU 2302 controls the phase component in accordance with thegenerated change amount (step S2313). The CPU 2302 generates an enhancedsignal by using the phase component having undergone the controlprocessing in step S2313 (step S2314).

Accordingly, the same effects as those of the first embodiment can beobtained. Note that the same applies to the second to 11th embodiments.The system implemented when the CPU executes the signal processingprogram for implementing the functions of the embodiments is alsoincorporated in the scope of the present invention.

While the present invention has been described with reference toexemplary embodiments, it is to be understood that the invention is notlimited to the disclosed exemplary embodiments. The scope of thefollowing claims is to be accorded the broadest interpretation so as toencompass all such modifications and equivalent structures andfunctions.

This application claims the benefit of Japanese Patent Application No.2012-259218 filed on Nov. 27, 2012, which is hereby incorporated byreference herein in its entirety.

What is claimed is:
 1. A signal processing apparatus characterized bycomprising: a transformer that transforms a mixed signal, in which afirst signal and a second signal coexist, into a phase component foreach frequency and one of an amplitude component and a power componentfor each frequency; a change amount generator that generates a changeamount of the phase component at a predetermined frequency by using aseries of data with a cross-correlation weaker than that of the phasecomponents and randomness lower than that of random numbers; a phasecontroller that controls the phase component by using the change amountprovided from said change amount generator; and an inverse transformerthat generates an enhanced signal by using the phase component havingundergone control processing by said phase controller.
 2. The signalprocessing apparatus according to claim 1, characterized in that saidchange amount generator generates a change amount by using a series ofdata based on the phase components derived by said transformer.
 3. Thesignal processing apparatus according to claim 2, characterized in thatsaid change amount generator generates a change amount by using a seriesof data obtained by inverting signs of the phase components derived bysaid transformer for at least alternate sample values.
 4. The signalprocessing apparatus according to claim 2, characterized in that saidchange amount generator generates a change amount by using a series ofdata obtained by shifting the phase components derived by saidtransformer by at least one sample.
 5. The signal processing apparatusaccording to claim 2, characterized in that said change amount generatorsets, as a change amount, a phase component at a position symmetrical toa position of an original phase component with respect to a position ofhalf the total number of samples in one frame.
 6. The signal processingapparatus according to claim 2, characterized in that said change amountgenerator sets, as a change amount, a series of data obtained byreplacing the phase components derived by said transformer in samples ofone frame.
 7. The signal processing apparatus according to claim 2,characterized in that said change amount generator obtains a correlationbetween adjacent samples of the phase components, and determines thechange amount so as to eliminate the obtained correlation.
 8. The signalprocessing apparatus according to claim 1, characterized in that saidchange amount generator uses, as the series of data, a plurality ofvalues of amplitude components at frequencies higher than thepredetermined frequency.
 9. The signal processing apparatus according toclaim 1, characterized in that said change amount generator uses, as theseries of data, a plurality of values obtained by collecting theamplitude components at the frequencies higher than the predeterminedfrequency in the frequency direction.
 10. The signal processingapparatus according to claim 1, characterized in that said change amountgenerator uses, as the series of data, a plurality of values ofamplitude components of specific frequencies along the time axis. 11.The signal processing apparatus according to claim 1, characterized inthat said change amount generator includes an amplitude holding unitthat holds the amplitude components at the specific frequencies, anduses, among the held amplitude components, an amplitude component at thepredetermined frequency as the change amount of the phase component. 12.The signal processing apparatus according to claim 1, characterized inthat said change amount generator inputs the phase component transformedby said transformer, and generates a change amount with a weakcorrelation with the phase component.
 13. The signal processingapparatus according to claim 1, characterized in that said phasecontroller performs one of replacement and rotation for the phasecomponent by using the change amount provided from said change amountgenerator.
 14. A signal processing method characterized by comprising:transforming a mixed signal, in which a first signal and a second signalcoexist, into a phase component for each frequency and one of anamplitude component and a power component for each frequency; generatinga change amount of the phase component at a predetermined frequency byusing a series of data with a cross-correlation weaker than that of thephase components and randomness lower than that of random numbers;controlling the phase component by using the change amount generated inthe generating the change amount; and generating an enhanced signal byusing the phase component having undergone control processing in thecontrolling.
 15. A non-transitory computer readable medium storing asignal processing program for causing a computer to execute a method,characterized by comprising: transforming a mixed signal, in which afirst signal and a second signal coexist, into a phase component foreach frequency and one of an amplitude component and a power componentfor each frequency; generating a change amount of the phase component ata predetermined frequency by using a series of data with across-correlation weaker than that of the phase components andrandomness lower than that of random numbers; controlling the phasecomponent by using the change amount generated in the generating thechange amount; and generating an enhanced signal by using the phasecomponent having undergone control processing in the controlling.